Sub-THz Polarization-Dependent Beam-Steerable Metasurface Antenna-in-Package for 6G Communication Constructed Using an LTCC Process
Article information
Abstract
In this paper, a polarization-dependent beam-steerable metasurface lens that is integrated with an antenna-in-package (AiP) using a single low-temperature co-fired ceramic (LTCC) process is proposed for sub-terahertz applications. The novelty of this work lies in the co-design of a voltage-controlled oscillator, a stacked patch antenna, and a reactive impedance surface within a compact AiP configuration, all fabricated through a single LTCC process. This integrated design not only simplifies fabrication and reduces interconnect loss but also compensates for bonding-induced mismatches. To overcome the high propagation loss in the W-band, a polarization-sensitive metasurface lens is employed, enabling significant gain enhancement and beam steering based on the antenna’s polarization. Notably, the three types of three- and five-layered unit cells used in this study feature a grid or cross-dipole slot positioned between the single/stacked patch on each outer side. In contrast to the conventional three-metal-layered process that allows for 180° phase coverage to ensure identical thickness, the proposed unit cell achieves full 360° phase coverage with a transmission loss of less than 3 dB. The measured peak gain of the 100 GHz AiP with a baseboard is 8.86 dBi. For the AiP with the metasurface lens, the beams in the boresight, +30°, and +40° directions at 100 GHz under x-axis polarization exhibit peak gains of 17.8, 16.2, and 16.0 dBi, respectively. For y-axis polarization, the beam in the boresight, −30°, and 20° directions achieve peak gains of 17.7, 16.6, and 16.5 dBi, respectively. Overall, the proposed scheme demonstrates promising beam steering for both types of polarization with a single fabrication process for W-band systems.
I. Introduction
The sub-terahertz (THz) frequency band is rapidly emerging as a candidate for sixth generation (6G) wireless communication, enabling ultra-high-speed data transmission by utilizing a high absolute bandwidth and increasing the frequency above the W-band. Currently, packaging technology that satisfies the market demand for miniaturization and low power consumption in millimeter-wave band fifth-generation (5G) wireless communication is being widely studied. Compared to 5G communication, the W-band signal has at least twice as many and three times smaller wavelengths. Moreover, considering that the loss tangent of a substrate increases with frequency, an ultra-compact and low-loss packaging solution is essential for 6G communication.
Packaging implementation methods can be divided into antenna-on-chip (AoC) and antenna-in-package (AiP), depending on the chip and antenna configuration. AoC involves the integration of a chip and an antenna on the same die through semiconductor chip fabrication. Owing to the chip size being limited to an area of several square millimeters, most AoCs are primarily used for the 200 GHz band and are not suitable for the W-band [1–5]. In contrast, AiP is a method in which the antenna and chip are manufactured separately and then packaged through additional bonding processes, such as gold studs, copper pillars, solder bumps, wire bonding, and wedge bonding. Since AiP technology typically exhibits higher area efficiency than the AoC approach, employing it to design antenna modules for the W-band can help achieve a high gain of 10 dB or more [6–8]. Nonetheless, despite its many advantages, AiP in the W-band still involves several significant challenges due to its high frequency (above 100 GHz).
Most state-of-the-art sub-THz and millimeter-wave AiPs focus primarily on antenna design, assuming an output impedance of 50 Ω for an integrated circuit (IC) [9, 10]. In general, both the IC and antenna impedance are designed for a return loss of 10 dB or more with respect to an impedance of 50 Ω. However, if the impedance is in the inductive/capacitive or low/high resistive region on the Smith chart, the renormalized return loss may decrease below 10 dB. Moreover, when designing a W-band AiP, the impedance mismatch effect of the bonding structure, which is not considered in the millimeter-wave band, may lead to a decline in return loss. For instance, wedge bonding is an interconnection method implemented on many packaging materials for cost-effective assembly. In this method, the insertion loss increases with the bonding length, and the bandwidth performance degrades owing to the impedance mismatch between the wedge and the IC pad. For return losses greater than 10 dB and insertion losses lower than 1.5 dB, a bonding length of approximately 0.02λ0 is required [11]. Therefore, the mismatch problem caused by the parasitic components of the wire can be addressed by maintaining a length of λg/2 for self-impedance matching during bonding [9]. In this context, molding encapsulation allows for a bonding length of 600 μm at 120 GHz, although it is accompanied by signal losses and AiP volume augmentation problems owing to the compound. Notably, in the absence of molding, a bonding length of 1.25 mm is required at 120 GHz. Additionally, a wedge-bonding package exists between the antenna and the power detector [12]. However, current approaches for solving such mismatch problems are impractical since they require a high level of accuracy under strict conditions. A more detailed analysis of this phenomenon is presented in Section II. To address these issues, a co-design comprising the IC, antenna, and bonding structures involved in packaging must be established to alleviate process confinement using a technique that does not involve molding or increasing the size.
Another challenge is the complexity of fabricating ultra-compact AiP modules suitable for the microscopic size of a 100 GHz antenna. In addition, a line width and spacing of 100 μm or less may be required to implement the 50 Ω transmission line using a microstrip line, stripline, or coplanar waveguide (CPW), implying that general printed circuit board (PCB) processes cannot be implemented. Furthermore, the short wavelength at sub-THz frequencies necessitates an approach that can increase radiation efficiency, owing to the surface wave and field leakage effects. High efficiency can be achieved by guiding the low-loss signal through a fine high-density pitch via an accurate arrangement. However, a PCB process with a minimum via diameter of 200 μm and a via pitch of 300 μm is unsuitable for the W-band. As a result, materials such as metal-clad polytetrafluoroethylene (PTFE) substrates, all-in-polymer multichip modules (MCM-Ps), liquid crystal polymers (LCPs), and polymer build-up on glass have been used to fabricate sub-THz AiPs [13–16]. While the PTFE process is attractive because of its low tanδ, it requires a high temperature and high pressure for lamination and thermal compression. Furthermore, AiPs fabricated using MCM-P offer the advantage of eliminating bond-wire interconnections, but exhibit high losses in transmission lines. Meanwhile, the LCP process can easily implement large-area patterns but has poor thermal stability. In an AiP configuration, where most of the antenna and IC are tightly attached, the heat of the IC is directly transferred to the antenna. A lack of thermal stability can result in critical reliability problems. The glass-based process exhibits a high process error of 18% or more under process conditions of 100 μm or less, which may result in problems with manufacturing stability and repeatability.
The low-temperature co-fired ceramic (LTCC) process, which addresses the limitations of the other processes while also satisfying the requirement for a via arrangement characterized by a high density level and fine pitch, is often used as a promising antenna packaging technique for the W-band to sub-THz frequencies [9, 17–19]. An important advantage of LTCC, which is a lamination process, is that it is characterized by various ceramic dielectric constants and stacks of metal layers with thickness units of at least 50 μm. Therefore, one of the main advantages of using LTCC for the lamination process is that it offers a flexible layer configuration with various dielectric constants and a thickness of 50 μm or more. Effectively, with LTCC, precise metal printing is possible while maintaining a minimum line width of 60 μm and a line spacing of 50 μm. In this study, a low dielectric-constant YT-6 ceramic (ɛr = 5.9, tanδ = 0.01 at 100 GHz) was employed to minimize dielectric losses of the antenna and the metasurface lens layout. Moreover, since ceramic material has a coefficient of thermal expansion identical to that of IC silicon die, the heat generated by the additional interconnection process would cause only minor errors.
The final challenge is the limited application area. Although high performance can be achieved by increasing the frequency, AiP research focuses on short-range communication systems owing to attenuation problems [19–21]. Until the current 5G era, antenna array structures with beam-steering functions achieved high gain and were extensively employed to build efficient communication systems. However, this complicated the feedline structure and increased the number of feedlines, thus emerging as another loss factor. In this context, for beam steering, a power amplifier (PA) and a phase shifter structure were required for each feed line. Generally, PAs have low output power and efficiency, while phase shifters exhibit high loss levels above the millimeter-wave bands. This implies that achieving high gains using array antennas above 100 GHz is limited by factors such as an increased RF chain and reduced monolithic-microwave-IC area utilization.
A metasurface lens is a structure that artificially changes the properties of radio waves. It is composed of unit cells that exhibit a spatial filter response. It is an essential technique for improving antenna performance despite constraints on the number of arrays due to the reasons outlined above, where distortion occurs during the embedding process in the final packaging stage. Owing to their low-profile characteristics, metasurfaces have been actively studied, even for application at frequencies above 100 GHz [22–26]. However, research on bulky dielectric lenses has demonstrated that they are not suitable for miniaturization. Moreover, most metasurface lenses have been designed to perform the singular function of increasing antenna gain. Even a beam-steerable lens operates only for a single type of incident wave polarization. To the best of our knowledge, we propose the first AiP design featuring a lens that can steer the beam using a selectable value while also increasing the gain based on the polarization of the antenna. Fig. 1 shows a schematic diagram of the lens integrated with the AiP module, whose application can be extended to long-range operations when a narrow beam is required. The lens jig was implemented using a dummy LTCC ceramic substrate, which was completely packaged in a single process.
Proposed W-band metasurface lens-integrated AiP topology.
Section II describes the design methodology for LTCC AiP and the co-design of the voltage-controlled oscillator (VCO). The LTCC unit-cell structure and lens design process are described in Section III. Section IV presents the measurement results of the AiP, with and without the metasurface lens. Finally, the results of a comparative analysis considering other lenses reported in earlier studies are discussed.
II. Antenna-in-Package Design
1. Co-design of the LTCC Antenna and CMOS VCO
Fig. 2(a) shows that the proposed AiP comprises an IC, a bonding structure, and an LTCC antenna. To operate as the primary signal source while also verifying performance, the IC is composed of a VCO generating a signal of 100 GHz, designed by implementing a 28-nm CMOS process. The widely used cross-coupled topology was adopted to meet the oscillation condition for generating a stable signal. The 100-GHz signal was extracted from a fundamental oscillation frequency of 50 GHz using a push–push technique, which extracted the output signal from the center tap of the resonant tank inductor. As is customary in general CMOS implementations, Fig. 2(a) shows that the output configuration of the chip includes a ground–signal–ground (GSG) pad for probing measurements and AiP integration. Fig. 2(b) shows the target-matching impedance range of the LTCC antenna based on the bonding inductance and ZVCO, including the capacitance effect of the pad. It is observed that the 10 dB return-loss matching circle normalized to 50 Ω is different from the impedance trajectory of the antenna. These must be practically matched, considering the packaging effect.
(a) Schematic of the AiP with the IC configured as a VCO and (b) impedance range of each node.
The AiP consists of a 100 GHz antenna bearing the structure of a well-proven 60 GHz LTCC parasitic stacked patch antenna [27] (Fig. 3). The parasitic patches introduced an additional resonance that significantly extended the bandwidth. The signal was directly fed to the driven patch by a straight via, given that signal loss during the via-to-line conversion process was not negligible and the ground vias overlapped at 150 μm intervals to minimize field leakage. The parameters of the antenna design are listed in Table 1.
LTCC antenna structure with parasitic patches for wideband operation: (a) 3D view and (b) top view.
Design parameters of the proposed LTCC antenna (unit: mm)
As shown in Fig. 4(a), the initial matching design exhibited dual impedance bandwidths of 85.9–107.4 GHz (22%) and 114.9–124 GHz (7.6%). The corresponding radiation beam patterns at each resonance frequency are depicted in Fig. 4(b). It is observed that at 100 GHz, the beam maintains a proper direction without noticeable distortion, but the maximum gain is relatively low—6.7 dBi. Furthermore, the surface current distribution illustrated in Fig. 4(c) indicates that electric coupling is dominant at 100 GHz. In such a scenario, unequal coupling strengths between the driven patch and adjacent parasitic patches result in suboptimal power transfer, which limits radiation efficiency and gain, even though the beam shape is preserved. At 120 GHz, as shown in Fig. 4(d), magnetic coupling dominates, and the beam splits symmetrically along the y-axis, leading to pattern degradation and a further reduction in gain, which declines to 5.8 dBi. To resolve these issues, the spacing between the driven and parasitic patches was carefully optimized to balance the coupling intensities. This coupling optimization enabled the achievement of improved gain without beam distortion. The final design achieved a wide impedance bandwidth of 89.2–111.2 GHz (22%) and an enhanced gain of 6.9 dBi at 100 GHz, which are suitable for accurate performance verification. Furthermore, the impedance of the optimized antenna was calculated to be 40.3–j·16 Ω at 100 GHz, which is within the target impedance range.
Simulation results of the proposed antenna: (a) reflection coefficient and (b) radiation patterns; current distributions on patches at (c) 100 GHz and (d) 120 GHz.
2. Analysis of Package Bonding and RIS
To couple the VCO to the antenna, a minimum bonding length of 0.05λ0 or less had to be maintained to ensure a wide bandwidth and to minimize RF signal loss. For ultra-compact and high-density integration, the same height of the cavity with a 200 μm-thick VCO was manufactured and stationed at the bottom of the antenna as an IC attachment (Fig. 5). A part of the ground via adjacent to the signal via was removed to minimize the distance of the cavity-to-signal via. When a cavity structure is added, a minimum module thickness of 600 μm must be ensured for stability. The antenna module size was set to 3 mm × 16 mm, which was increased in a direction independent of the polarization of the antenna for insignificant performance variations. The bottom side of the cavity was plated for the ground connection of the chip, bearing a size of 1.5 mm × 1.5 mm, with a margin to reduce process complexity. The dimensions of the AiP design are listed in Table 2.
Design parameters: (a) 3D view of the AiP, (b) RIS structure of Layers 3 to 5, (c) diagram of the cavity, and (d) wedge bonding topology.
Design parameters of the proposed LTCC antenna with RIS (unit: mm)
Several solutions have been proposed to address the impedance mismatch problem caused by the local bonding area, such as the self-matching method and the reactive impedance surface (RIS) method. In self-matching, a bonding length of 1.5 mm (λg/2) would increase the package volume. However, in the LTCC process, metals can be stacked in units of 50 μm. Therefore, up to three layers of RIS were inserted between the directly fed patch in Layer 2 and the ground in Layer 6, without taking recourse to any additional stacking. A RIS consists of a periodic arrangement of square patches with a minimum pattern width of 60 μm and a gap of 50 μm. The parasitic inductances of the wedge bondwires can be compensated for by conjugating the antenna input impedance using a multilayer RIS structure. Fig. 6 depicts a comparison of the antenna performance achieved using the self-matching technique and different numbers of RIS structure layers. Compared to self-matching, increasing the number of RIS layers from 1 to 3 improves the impedance matching bandwidth by 4.8%, 7.5%, 8.8%, and 17.7%, respectively. The simulation results indicate that the three RIS layers improved the impedance bandwidth by a factor of 3.67. Furthermore, the gain beam pattern results indicated that the bandwidth enhancement resulting from the addition of RIS structures was not due to signal loss, with the gain
Comparison of the results for (a) impedance matching and (b) radiation pattern using RIS and self-matching.
As shown in Fig. 7(a), the improved impedance bandwidth resulted in a small performance variation in the cavity structure fabrication for the VCO chip insertion and wedge-bonding connection errors. Notably, it is difficult to guarantee precise wedge-bonding dimensions and an even flat cavity using the LTCC process. Therefore, the height of the chip protrusion, wedge-bonding height, and length variations were accounted for in the simulation by varying Hdiff, Hwedge, and Lwedge. Considering the 10 dB return loss obtained through optimized bonding, along with the maximum impedance bandwidth, acceptable process tolerances for Lwedge, Hdiff, and Hwedge were considered to be ±100 μm, ±200 μm, and ±125 μm, respectively. Notably, the tolerance ranges being sufficiently larger than the process error by several tens of μm confirms that the proposed AiP can be fabricated practically. Fig. 7(b) shows the variation in the realized gain over the acceptable error range. The results indicate a maximum gain of 6.18 dBi and a gain variation of 0.72 dB within the tolerance range at 100 GHz. Furthermore, in Fig. 7(c), the optimized wedge bonding for the AiP is represented as a modulated π-model equivalent circuit [26, 27]. A comparison of the electromagnetic simulation results with those obtained using the equivalent circuit composed of a self-inductance of 77 pH and parasitics shows that the two impedance trajectories on the Smith chart are almost identical. The transmission coefficient and phase shift of wedge bonding, extracted using the equivalent circuit model, are shown in Fig. 7(d). The loss is 1.54 dB at 100 GHz, as the change in loss is 0.86 dB over the 90–110 GHz frequency range.
Variations in AiP: (a) reflection coefficient and (b) radiation pattern performance at 100 GHz with manufacturing errors in the wedge bonding and cavity process; (c) equivalent circuit, (d) insertion loss and phase of optimized bonding.
III. Metasurface Lens Package Design
1. Unit Cell Selection
The unit cells considered in this study comprise stacked patches that are mutually coupled to a grid or a cross-dipole slot, which contributes to improving the polarization isolation of antennas. At sub-wavelengths, the capacitance component of the patch and the inductance component of the grid were observed to be dominant. Notably, the cross aperture of the dipole acts as a parallel LC resonator. Therefore, the unit cell exhibit a bandpass response, with the order of the corresponding filter increasing with the number of metal layers.
Notably, the phase shift value of the unit cell changed rapidly with frequency, thus securing a broader phase modulation range. Fig. 8(a) depicts a unit cell structure designed using the conventional PCB process. Notably, a narrow phase range and high insertion loss of the PCB-based unit cell result from the insertion of bondply material, with the minimum pattern size being 100 μm for both polarizations [30]. In this study, each unit cell has the same thickness of 300 μm (0.1λ0) and a size of 600 μm × 600 μm (0.2λ0). The unit cell structure consists of a grid or cross-dipole aperture between one or two stacked patches (Fig. 8(b)–8(d)). Seven metal layers are required to design the lens, with the proposed unit cells arranged based on the thickness of each layer. The S2112 parameter was neglected because each proposed LTCC unit cell exhibited a high polarization characteristic of 50 dB or more. Furthermore, unit cells with independent phase-shift values for each polarization were required for different beam-steering features with respect to antenna polarization. Therefore, the patch, grid, and cross-dipole slot constituting the unit cell structure had independent lengths along the x- and y-axes. A total of 144 unit cells were employed to achieve a phase-tunable range of 360° in units of 30° for the two linear polarizations. After 72 unit cells were defined, the remainder were automatically determined since the unit cell characteristics were symmetrical along the two orthogonal axes.
Unit cell structure: (a) a conventional PCB unit cell, (b) the proposed three-metal-layer structure, (b) the five-metal-layer structure with a grid, and (d) a cross-dipole slot LTCC unit cell.
As shown in Fig. 9(a) and 9(b), the phase-tunable range when using only three metal unit cells is 180° for each polarization, whereas the unit cells with five metal layers achieve a range of 360° for both types of polarization. Notably, the figures depict the phase-tunable range for only one type of polarization. Table 3 summarizes the design parameters of the unit cells shown in Fig. 9. Furthermore, Fig. 9 shows that when the x-axis polarized wave (X-pol) is incident, it is uniformly distributed over a 360° range in units of 30°, as indicated by the dotted line, whereas the solid line shows that the phase distribution is confined to a specific value for the y-axis polarized incident wave (Y-pol). The inserftion loss based on the frequency of each unit cell is shown in Fig. 9(c) and 9(d). Although unit cells satisfying an insertion loss of less than 3 dB were selected because of high signal attenuation loss in the W-band, most cells exhibited insertion losses less than 1.5 dB at 100 GHz. Notably, when multiple unit cells have the same phase-shift value, a unit cell with less insertion loss is adopted during the final selection process. Moreover, if unit cells having the same 360° phase-tunable range consider only one polarization scenario, a sufficiently low insertion loss condition will be realized.
Phase-tunable range of the selected unit cell: (a) three-metal-layer unit cell, (b) five-metal-layer unit cell; insertion loss of the (c) three-metal-layer unit cell, (d) five-metal-layer.
Design parameters of the proposed LTCC unit cell
2. Dual Polarized Lens Design Process
The first step in the lens design process was to capture the phase of the fields radiating from the AiP. The effect of the AiP radiation field due to the LTCC jig and the top metal of the baseboard was also accounted for in Fig. 10(a). Fig. 10(b) shows the tapering, spill-over, and feed efficiency of the lens at each focal length. As shown in Fig. 10(c), lens placement at short focal lengths, where the amplitude change of the captured field is discrete, should be avoided to achieve a unit cell zone. Therefore, in the proposed metasurface AiP, the lens was placed at a focal length of 10 mm, given its compact design and high efficiency. A total of 784 unit cells were used to configure the LTCC metasurface, with 28 unit cells—0.6 mm in size— arranged along each axis. A final lens f/D value of 0.59 was selected for zone formation, thereby achieving a compact system.
Unit cell structure: (a) phase capture diagram considering the field effect of the metasurface lens jig and baseboard, (b) lens efficiency, and (c) phase distribution at the center of the lens.
Fig. 11 illustrates the lens design process using the selected unit cells. Given that the captured phase distribution was symmetric to the y-axis, the lens design sequence for half the area is shown in Fig. 11(a). The captured phase was compensated for using a specific constant value to convert the spherical wave into a plane wave. In addition, a specific phase value was determined to satisfy the scenario in which unit cells with low insertion loss exhibited the greatest level of dispersion in the final stage.
Dual polarization metasurface lens design process: (a) captured phase distribution of the AiP with a focal length of 1 cm (b) phase compensation and phase offset provision stage, (c) unit cell matching, (d) unit cell zone formation for the performance improvement technique, and (e) final unit cell distribution diagram for X-pol and Y-pol.
The antenna phased array theory involves a phase offset process in which a uniform phase difference must be applied to each array of unit cells located at regular intervals along the y-axis to steer the beam. When the desired beam steering angle is determined, the offset value can be easily calculated from the array factor (Fig. 11(b)). Subsequently, the unit cells corresponding to the phase values at each position can be matched. As shown in Fig. 11(c), if unit cells corresponding to the phase of each position are accurately arranged, the performance of the lens degrades [31]. However, this problem can be solved by zone formation, which refers to the grouping of adjacent unit cells with similar phase values. Another phase distribution (Fig. 11(d)) can be obtained by repeating the same process for other types of polarization. For convenience, the unit cell distribution for the two polarizations are combined into one diagram in Fig. 11(e).
IV. Fabrication and Measurement Results
1. Fabrication
Fig. 12 shows the structural view and photograph of the bottom side of the LTCC antenna, observed under a microscope, depicting its wedge bonding with the VCO. Using the LTCC process, the cavity was machined such that the average distance from the signal through the edge to the cavity, as indicated by the cavity−via parameter, was 57.4 μm. In addition, plating for the electrical connection of the chip was achieved without having to account for rigorous limitation conditions. Since the cavity area was sufficiently wider than the chip area, it bonded to the bottom of the cavity, providing a ground connection. For DC bias bonding, a rectangular pad was fabricated, accounting for the wedge tail. The pad was electrically connected to the circular pad located at the bottom right through a bias line inside the LTCC module, which was then solder ball-bonded to the baseboard and directly attached to the power supply. Fig. 13 shows the three fabricated lenses with steering angles of (0, 0), (30, 30), and (40, −20) for X-pol and Y-pol. The thickness of each lens module for the seven metal layers was 300 μm, while the minimum printed pattern width and gap were 60 and 50 μm, respectively. Fig. 14 depicts the solder-ball bonding between the AiP and the baseboard, with the voltage supplied through a DC connection. Notably, the baseboard was fabricated from a 3 mm × 3 mm FR4 substrate. The metasurface lens was placed at a focal length through the LTCC jig, which aided in reducing the sidelobes caused by baseboard interconnection.
(a) Structure view and (b) photograph of the bottom side of the LTCC antenna wedge-bonded with the VCO.
Photograph of the fabricated lens with beam steering angle of (a) (0,0) (b) (30, −30), and (c) (40, −20) for X-pol and Y-pol.
Photograph of the test board combined with (a) only AiP and (b) AiP with the metasurface lens.
2. AiP Measurement
To verify the performance of the LTCC antenna before VCO attachment, the reflection coefficient of the W-band was measured using a FormFactor i110 150 μm pitch GSG probe. A transmission line with a minimum length of 50 μm was attached to the bottom side of the fabricated antenna, surrounded by ground metal spaced 50 μm apart to reduce signal loss. The S11 of the antenna was verified by attaching the flipped antenna onto Styrofoam, which had a dielectric constant identical to that of air and a thickness of 5 mm to satisfy the far-field condition. Return loss was measured through single-port measurement using the W-band extender module of a vector network analyzer. Fig. 15 plots the simulated and measured reflection coefficients, with the measured impedance bandwidth being 23 GHz (83–106 GHz). Notably, the transmitting AiP operated the VCO by applying a DC voltage to generate a signal, which was subsequently radiated through the LTCC antenna using bonding. Measurements were conducted at 100 GHz by tuning the frequency to account for the VCO design conditions. Maintaining a distance of 6 cm from the receiver, a 10–7025 Aerowave WR10 horn antenna was employed as the receiving antenna. The absolute power of the downconverted signal was measured using a Keysight E4448A spectrum analyzer through a Keysight 11970 W harmonic mixer (Fig. 16). The AiP gain amplitude was normalized to a peak gain of 9.2 dBi by calculating the free-space loss, down-conversion loss by the mixer, and cable loss in the W-band. Signals in the 90–110 GHz band generated through the Keysight E8275D signal generator were delivered to the Formfactor i110 100 μm pitch GSG probe, electrically connected via the CPW line, which is nearly lossless. The power was measured with the same mixer and spectrum analyzer used in the AiP setup. Various loss factors, such as those of the mixer, cable, and probe values, over the 90–110 GHz band were calibrated using a power meter to determine the absolute power at the harmonic mixer and signal generator. Fig. 17 presents the measured results of the AiP beam pattern at 100 GHz, obtained through a ±90° rotation using an automatically controlled turntable. The measured peak gain and sidelobe level were 8.86 dBi and 3.24 dB, respectively, indicating good agreement with the simulation results.
Reflection coefficient of the simulated and measured probing antenna results.
Gain and radiation pattern measurement setup for the AiP and the metasurface lens.
Measurement of the AiP beam pattern at 100 GHz.
3. Metasurface Lens Measurement
Two of the three types of polarization-dependent metasurface lenses were designed to have beam steering angles of ±0° and ±30° for each type of polarization, where the sign indicates the corresponding beam steering direction for X-pol and Y-pol. To verify that the steering angles would differ for each polarization type, we manufactured and measured a lens that bent the peak beam at +40° for X-pol and -25° for Y-pol. Lens gain calculation was easily performed by considering the difference between the received power values in the spectrum analyzer with and without lenses using the measurement setup illustrated in Fig. 16. The results showed that the beamwidth decreased as the gain increased. Notably, owing to the output limit of the VCO, the measurements were performed only at an angle around the peak gain.
The simulation and measurement results for the three metasurface types at 100 GHz are presented in Fig. 18, indicating good agreement. For the lens that enhanced the gain in the boresight direction for both types of polarization, the simulation and measurement gains for X-pol were 18.5 and 17.8 dBi, respectively, while the simulation and measurement gains for Y-pol were 18.9 and 17.7 dBi, respectively. For the lens designed to steer the beam at 30° for both types of polarization, the simulation and measurement gains were correspondingly 17.0 and 16.2 dBi for the X-pol incident wave, and 17.1 and 16.6 dBi for the Y-pol incident wave, respectively. Similarly, the lens designed to steer the beam by +40° and -25° for X-pol and Y-pol operated as intended. The simulation and measurement gains were 16.3 and 16.0 dBi for the X-pol incident wave, and 17.2 and 16.5 dBi for the other types of polarization. Fig. 19 shows the boresight gain of the metasurface AiP with regard to the frequency. Notably, the frequency range of the measurement results was limited by the oscillation frequency range of the fabricated VCO. Fig. 19 shows that the X-pol and Y-pol 3-dB gain bandwidths are 98–105 GHz and 98–104 GHz, respectively, while the maximum aperture efficiencies are 15.2% and 15.1%, respectively.
Simulation and measurement results of the metasurface lens: (a, b, c) pertain to lens that steer a beam at 0, +30, and +40 degrees, respectively, for the X-pol incident wave (E-plane), while (d, e, f) pertain to lens that steer a beam at 0, −30, and −20 degrees, respectively, for the Y-pol incident wave (H-plane).
Simulated and measured gain and corresponding efficiency of the metasurface AiP.
4. Analysis and Discussion
The typical processes optimized for lens fabrication are PCB and LTCC. Table 4 presents a performance comparison of the proposed lens and state-of-the-art metasurface lenses that operate in W- and D-bands [21–24]. Prior studies on metasurface lenses for the W- and D-bands only considered combinations with standard horn antennas, without accounting for packaging using RF modules. In contrast, this paper proposes a metasurface lens capable of independently steering a beam in dual polarization, while also being integrated with AiP through the same process employed for beam steering. Lens is composed of seven metal layers with a thickness of 300 μm and a phase modulation range of 360°, while also achieving the smallest f/D value of 0.59. Compared to the PCB lens capable of beam steering at an operating frequency of 145 GHz, more than the current metal stack is possible, demonstrating that a sufficient improvement in gain can be achieved for the source antenna using LTCC. Furthermore, compared to the beam-steerable PCB lens operating at 145 GHz [22], the LTCC process employed in the present study can be used to design a small unit cell with a via-less structure using more than twice the number of metal layers. In contrast, although the PCB lens offers the advantages of a short focal length and beam steering, it is limited to one linear polarization (LP). Meanwhile, another beam steering approach proposed in a previous study combined multiple source antennas with a Huygens metasurface lens and then fed the antenna at an arbitrary location [25]. This approach is more efficient than a phased array antenna in a beam steering system, but it requires multiple feed lines. Furthermore, a beam steering angle of ±12° is possible when the feed position is moved 6 mm from the center of the lens. Most lenses reported in previous studies have been designed based on a standard horn antenna in the W- and D-bands, whereas the PCB metasurface operating at 130 GHz was designed based on AiP, achieving a simulation gain of 8.4 dBi [26]. Notably, the unit cell selection tasks considered in this study to ensure independent phase shift values for scenarios with dual LP modes are more stringent than those associated with previously studied unit cell designs, which only serve to increase the boresight gain or steer the beam, with gain enhancements for a single LP. Moreover, since a high insertion loss is inevitable owing to the operating frequency, the proposed unit cell can be selected to satisfy an insertion loss of less than 3 dB, as desired in most dual LP scenarios.
Performance comparison table
V. Conclusion
This paper proposes a full LTCC package incorporating a metasurface lens for beam control in two polarizations, using a novel AiP polarization function in the W-band. The bandpass spatial filter response of the unit cell was improved by stacking multiple layers and fine metal into a low-profile flat lens. By achieving a phase-tunable range of 360°, we built a system capable of steering the beam up to ±45° based on the polarization of the incident waves. An AiP-based lens with a beam steering angle of nearly 90° was designed, achieving a 1.6 dBi decrease in the peak gain from 17.8 to 16.2 dBi for the X-pol incident wave, with a focal length of 10 mm and aperture size of 16.8 mm × 16.8 mm. Furthermore, a wedge-bonded antenna co-designed with a VCO by employing RIS structure ensured the achievement of impedance-matching characteristics that were insensitive to the conditions of the interconnection process. Overall, in this paper, a promising sub-THz system comprising a packaged antenna and a metasurface lens capable of beam steering, depending on the type of AiP polarization, was fabricated through a single LTCC process.
Notes
This work was supported by the 2025 Research Fund of the University of Seoul.
References
Biography
Seongwoog Oh, https://orcid.org/0000-0002-7013-0586 received the B.S. degree in electrical engineering and computer science from the Gwangju Institute of Science and Technology College, Gwangju, South Korea, in 2016, and the M.S. and Ph.D. degrees in electrical engineering from Seoul National University, Seoul, South Korea, in 2018 and 2023, respectively. From 2023 to 2024, he was with the TICS Research Group, Department of Electrical and Computer Engineering, University of California San Diego, as a postdoctoral researcher. From 2024 to 2025, he was a faculty member at the Department of Electronics Convergence Engineering, Kwangwoon University in South Korea. He is currently an assistant professor with the School of Electrical and Computer Engineering, University of Seoul in South Korea. His research interests include the design of RF/millimeter-wave integrated circuits, antenna-on-package systems for 5G/6G communication, nuclear fusion antennas, and microwave bio-applications. He was a recipient of the 2019 IEEE MTT Seoul Chapter Best Paper Award, the 2022 IEEE AP-S Student Paper Competition Honorable Mention, and the 2022 IEEE Antennas and Propagation Society Fellowship.
Yeongmyeong Park, https://orcid.org/0000-0002-3737-5266 received her B.S. degree in physics and electronic and electrical engineering from Chung-Ang University, Seoul, South Korea, in 2019, and is currently working toward her M.S. degree at Seoul National University, South Korea. Her current research interests include the design of 5G mobile communication system antennas and metasurface lenses.
Jungsuek Oh, https://orcid.org/0000-0002-2156-4927 received his B.S. and M.S. degrees from Seoul National University, South Korea, in 2002 and 2007, respectively, and his Ph.D. degree from the University of Michigan at Ann Arbor in 2012. From 2007 to 2008, he was a hardware research engineer at Korea Telecom, working on the development of flexible RF devices. In 2012, he was a postdoctoral research fellow in the Radiation Laboratory at the University of Michigan. From 2013 to 2014, he was a staff RF engineer with Samsung Research America, Dallas, working as a project leader for the 5G/millimeter-wave antenna system. From 2015 to 2018, he was a faculty member in the Department of Electronic Engineering at Inha University in South Korea. He is currently an associate professor at the School of Electrical and Computer Engineering, Seoul National University, South Korea. His research areas include mmWave beam focusing/shaping techniques, antenna miniaturization for integrated systems, and radio propagation modeling for indoor scenarios. He is the recipient of the 2011 Rackham Predoctoral Fellowship Award from the University of Michigan. He has published more than 50 technical journal and conference papers, and has served as a technical reviewer for IEEE Transactions on Antennas and Propagation, IEEE Antenna and Wireless Propagation Letters, and so on. He has served as a TPC member and as a session chair for the IEEE AP-S/USNC-URSI and ISAP. He has been a senior member of IEEE since 2017.