I. Introduction
The rapid development of satellite communications, coupled with the limited spectrum resources in the L and S bands, has significantly increased the demand for millimeter-wave circularly-polarized (CP) antennas [1]. To meet the requirements of high data rates and stable communication services, satellite communication systems require antennas with enhanced bandwidth and stable high realized gain. CP arrays are commonly employed to achieve wideband high-gain radiation with circular polarization. Several designs with high aperture efficiency in a wide band have been reported [2–4]. However, these designs often incorporate complex feeding networks, which can introduce significant losses in the millimeter-wave band.
Fabry-Perot (FP) antennas offer the advantage of high gain with simple feeding structures, making them effective in avoiding transmission losses [5, 6]. Moreover, the literature in the past decade has presented FP antennas capable of achieving circular polarization [7]. However, in some cases, linearly-polarized (LP) radiators are preferred as the radiation source instead of CP ones, and linear-to-circular polarization converters are commonly employed [8]. These converters come in two types: reflection and transmission [9, 10]. Among these, the transmission type has been utilized in the design of CP FP antennas [11]. Additionally, polarization converters can be integrated into partially reflective surfaces (PRSs) to achieve compact designs [12–14]. For instance, a receiver-transmitter metasurface was proposed in [12], in which the unit cell consists of two patches acting as the receiver and the transmitter simultaneously, functioning as a circular polarizer and a PRS. However, the design in [12] exhibits limited impedance bandwidth and significant gain attenuation within the band.
The limited bandwidth and gain attenuation mentioned above can be attributed to the decrease in the reflection phase of the PRS as the frequency increases. This change in the reflection phase affects the resonant condition of the Fabry-Perot cavity (FPC), since the cavity thickness remains constant. To address this issue, several improved PRS designs with gentle and even positive reflection phase gradients have been reported. These designs ensure that the FPC resonant condition is met within a wide band [15] or even within dual bands [16]. Furthermore, a CP design based on a PRS with a gentle reflection phase gradient has also been proposed [17]. However, such a FP antenna typically adopts a CP radiation source, which often involves a complex feeding network.
It is anticipated that a PRS can be designed to simultaneously have a positive reflection phase gradient and function as a circular polarizer. In this letter, we propose a receiver-transmitter PRS that possesses both features and utilizes them in a CP FP antenna with a LP radiation source. On the one hand, the dual resonances resulting from the source radiator and the specific reflection phase contribute to an improved impedance bandwidth of 28.6–30.0 GHz for the final FP antenna. On the other hand, however, by employing polarization conversion and the specific reflection phase, the presented antenna achieves circular polarization with an axial ratio (AR) bandwidth of 28.6–29.6 GHz and a right-hand circular polarization (RHCP) gain fluctuation of less than 0.6 dB.
II. Antenna Design and Analysis
1. Antenna Geometry and Operation Principle
The geometry of the presented FP antenna is illustrated in Fig. 1. The antenna comprises two main components: the receiver-transmitter PRS and the LP radiation source. These two parts are separated by an air gap with a height of H0 = 5.0 mm. The PRS is designed with a periodic arrangement of 5 × 5 unit cells. The LP radiation source utilizes a substrate-integrated waveguide (SIW) cavity-backed slot, with an additional feeding slot etched on the bottom surface of the cavity. The SIW cavity is fed by a printed ridge gap waveguide (PRGW), which is chosen for its relatively low transmission loss in the millimeter-wave band.
The FP antenna has the advantages of its enhanced bandwidth, circular polarization, and stable, high realized gain, which can be explained from the following three aspects. Firstly, referring to [18], the resonant condition of the FPC can be expressed as
where f represents the operating frequency and ϕGND and ϕPRS are the reflection phases of the ground and the PRS, respectively. It is observed that a PRS with a positive gradient in the reflection phase (ϕPRS) can counteract the frequency-dependent effects on the resonance condition. Consequently, the design of a radiation source with dual resonances and the incorporation of a PRS with a positive reflection phase gradient enable the realization of enhanced bandwidth.
Secondly, the directivity of the FP antenna can be calculated using the following equation:
Here, Ds represents the directivity of the radiation source, while RPRS denotes the reflection coefficients of the PRS. It is evident from Eq. (2) that the final realized gain of the FP antenna is determined by both the radiation source and the PRS. The radiation source, with its low gain fluctuation, contributes to the stable performance of the antenna. On the other hand, the PRS, with its gentle reflection magnitude, further enhances the stability and high realized gain of the antenna. By carefully designing both the radiation source and the PRS, the FP antenna achieves a stable and high realized gain across the desired frequency band.
Finally, it should be noted that the FP antenna benefits from the integration of the linear-to-circular polarization conversion function within the PRS. This integration enables the FP antenna to generate CP radiation across the enhanced bandwidth.
2. Receiver-Transmitter PRS
Based on the aforementioned analysis, we propose a receiver-transmitter PRS based on a SIW cavity. Fig. 2(a) illustrates the structure of the PRS, which consists of two bow-tie-shaped slots etched on the top layer of the cavity at oblique angles of ±45°. The lengths of these slots are denoted as Ls1 and Ls2. These SIW cavity-backed crossed slots function as CP transmitters, similar to the crossed slot antenna design presented in [19]. On the bottom layer of the cavity, dual rectangle slots with identical lengths and widths are etched. These SIW cavity-backed dual slots serve as an LP receiver, which is an evolution of the design reported in [20]. The substrate used in this design is F4BM, with a relative permittivity of 3.5 and a thickness of t1 = 1.0 mm. The unit cell of the PRS exhibits the following physical parameters: Dc = 4.0 mm, Ds0 = 0.7 mm, Ds1 = Ds2 = 0.4 mm, Dv = 0.4 mm, Ls0 = 3.0 mm, Ls1 = 3.7 mm, Ls2 = 3.0 mm, Rv = 0.15 mm, Ws0 = Ws1 = 1.0 mm, and Ws2 = 0.8 mm.
Fig. 2(b) illustrates the simulation model of the unit cell for the receiver-transmitter PRS. The reflection and transmission coefficients are investigated specifically for x polarization. According to [21], the equivalent circuit model (ECM) can be established, as shown in Fig. 2(c). Detailed values of these equivalent components are: L1 = 1.125 nH, L2 = 0.0785 nH, L3 = 0.23 nH, C2 = 0.18 pF, and C3 = 0.18 pF. The impedance of the two ports of the ECM is Z0 = 377 Ω, and the dielectric substrate is modeled as a transmission line with its impedance of
Z 1 = Z 0 / ɛ r = 201 Ω .
Fig. 3 presents the simulated and calculated S-parameters of the PRS unit cell. In Fig. 3(a), it can be observed that the PRS exhibits a simulated varying reflection magnitude ranging from 0.68 to 0.75 within the frequency range of 28.0 to 30.0 GHz. This indicates that approximately half (46.25% to 56.25%) of the incident waves are reflected, while the remaining portion is transmitted. It is evident that the designed PRS effectively performs its partially reflective function. The calculated results also show dual resonances and partially reflective features. Furthermore, in contrast to traditional PRS designs, in which the reflection phase experiences a significant drop with increasing frequency, based on the simulated and calculated results, Fig. 3(b) demonstrates that the proposed PRS possesses a positive reflection phase gradient. This characteristic holds significant potential for achieving favorable FPC resonance and ensuring stable realized gain of the FP antenna throughout an expanded operating bandwidth.
Additionally, Fig. 3(a) demonstrates that the transmitted waves are separated into two components: the x- and y-polarized components (denoted as Txx and Tyx, respectively). Remarkably, within the frequency range of 28.0 to 30.0 GHz, these two components exhibit nearly identical magnitudes, with a difference of less than 0.1. Moreover, as depicted in Fig. 3(b), the x- and y-polarized transmitted waves maintain a phase difference of approximately 90° across this frequency range. The nearly equal magnitudes and orthogonal phase relationship of the transmitted x- and y-polarized waves within the specified frequency range allows the FP antenna to generate circularly polarized radiation in the desired frequency band.
Based on the simulation-based analysis conducted, the designed receiver-transmitter periodic structure demonstrates significant potential for utilization as a PRS within a bandwidth-enhanced and stable-gain FP antenna. The structure exhibits dual functionality, simultaneously serving as a PRS with a positive reflection phase gradient and a circular polarizer.
3. LP Radiation Source Fed by PRGW
The radiation source of the proposed antenna is a PRGW-fed slot antenna, as illustrated in Fig. 4. It utilizes an SIW cavity to facilitate dual resonances. The top surface of the SIW cavity features a rectangular slot, while a feeding slot is etched on the bottom surface. The SIW cavity is coupled to the slot and radiates through the radiation slot. To minimize dielectric losses, a PRGW feeding structure is employed. The electromagnetic parameters of the substrates used in the design are depicted in Fig. 4(a). The LP radiation source possesses the following specific physical parameters: Dp = 0.4 mm, L1 = 1.5 mm, L2 = 7.2 mm, Lc = 4.0 mm, Lfs = 2.8 mm, Lrs = 3.6 mm, Rp = 0.15 mm, t1 = 1.0 mm, t2 = 0.508 mm, t3 = 0.762 mm, W1 = 1.6 mm, W2 = 1.5 mm, Wc = 3.6 mm, Wfs = 0.1 mm, and Wrs = 0.2 mm.
The unit cell geometry of the PRGW is depicted in Fig. 5. It consists of a periodically arranged mushroom-shaped structure with a metal cover, which serves to achieve an electromagnetic band-gap function. The PRGW line and the metal cover enable the support of quasi-TEM modes. Additionally, Fig. 5 illustrates the dispersion diagram of the PRGW, revealing the presence of a band gap spanning the frequency range of 27.0 to 34.5 GHz.
The SIW cavity supports dual resonant modes, namely TE110 and TE210, as depicted in Fig. 6. The H-field distribution on the top surface of the cavity indicates that the TE110 mode is supported at 28.5 GHz, while the TE210 mode is supported at 29.5 GHz. It is anticipated that by optimizing the cavity size and the feeding structure, the source antenna can achieve dual resonances. Fig. 7 illustrates the reflection coefficients of the source antenna, revealing the presence of dual resonances. Furthermore, the S11 parameter remains below −10 dB from 28.5 to 30.8 GHz, indicating good impedance matching. The broadside gain of the source antenna is also shown in Fig. 7, demonstrating a low gain fluctuation across the aforementioned frequency range. This characteristic contributes to the stable-gain performance of the final FP antenna.
III. Simulated and Measured Results
The final CP FP antenna was fabricated based on the above analysis. Fig. 8 showcases both the individual antenna components before assembly and the assembled structure. After being assembled, the fabricated prototype was subjected to reflection and radiation measurements.
Fig. 9(a) presents the reflection curves of the FP antenna, revealing an impedance bandwidth of −10 dB from 28.6 to 30.0 GHz. The 3-dB AR bandwidth, as depicted in Fig. 9(b), is slightly narrower than the impedance bandwidth, spanning from 28.6 to 29.6 GHz. Fig. 9(c) compares the simulated and measured realized gains of the final antenna. Within the afore-mentioned bandwidth, the presented antenna achieves a peak gain of 12.1 dBi with a gain fluctuation of less than 0.6 dB. Notably, the measured broadside gain shows an enhancement of approximately 7.0 dB compared to the simulated results of the LP source antenna (as shown in Fig. 7) within the operating band. These results confirm that the designed FP antenna possesses the desired features of enhanced bandwidth, circular polarization, and a high stable gain.
It is worth noting that, as the frequency increases, the radiation aperture increases. As plotted in Fig. 9(a), although the reflection coefficient drops after the operation frequency passes 29.8 GHz, the increased radiation aperture can neutralize the increase in the return loss to some extent. Therefore, the total realized gain is able to maintain stability until the frequency reaches 30.5 GHz. On the other hand, it is worthwhile to note that the black and red gain curves plotted in Fig. 9(c) present the total gain but not the RHCP gain. For comparison, the RHCP gain curves are also presented in Fig. 9(c). It can be seen that the RHCP gain also drops rapidly when the operation frequency passes 30.0 GHz, which agrees well with the AR curves shown in Fig. 9(b).
Furthermore, Fig. 10 displays the simulated and measured normalized radiation patterns of the final FP antenna. It should be noted that, due to experimental limitations, only far-field results in the upper half space were measured. As predicted, the antenna exhibited boresight RHCP patterns. Additionally, the measurement results demonstrate low sidelobe levels and low cross-polarization for the final FP antenna.
Overall, the measured results validate the anticipated advantages of the designed FP antenna, including its enhanced bandwidth, circular polarization, high stable gain, and favorable radiation characteristics.
IV. Conclusion
The proposed stable-gain millimeter-wave CP FP antenna, based on a circular polarizer-integrated PRS with a positive reflective phase gradient, offers an enhanced bandwidth, circular polarization, and high stable gain. These features make it suitable for satellite communication and 5G millimeter-wave wireless communication systems. The experimental results validate the performance and effectiveness of the designed antenna, paving the way for further advancements in millimeter-wave antenna technology.