Hybrid Topology with Reconfigurable Rectifier for Enable CC and CV Output Characteristics in Wireless Power Transfer Systems

Article information

J. Electromagn. Eng. Sci. 2024;24(3):285-293

Publication date (electronic) : 2024 May 31

doi : https://doi.org/10.26866/jees.2024.3.r.229

Haibing Wen^1,, Jiayuan Li¹, Jiadong Yang¹, Peng Wang¹, Kehan Zhang², Lei Yang¹, Yaopeng Zhao¹, Xiangqian Tong¹

¹School of Electrical Engineering, Xi’an University of Technology, Xi’an, China

²School of Marine Science and Technology, Northwestern Polytechnical University, Xi’an, China

^*Corresponding Author: Haibing Wen (e-mail: wenhaibing@xaut.edu.cn)

Received 2023 July 5; Revised 2023 September 4; Accepted 2023 December 8.

Abstract

I. Introduction

Wireless power transfer (WPT) technology has the potential to effectively enhance the freedom and security of a power supply approach. It has been widely used in electric vehicles, rail transit, implantable medical devices, industrial equipment, consumer electronics, power system monitoring equipment, and other fields [1–8]. In general, lithium-ion batteries have been chosen as the power source for several types of electrical equipment. To enhance the safety and efficiency of battery charging, a typical charging process can be divided into two stages—constant current (CC) charging mode and constant voltage (CV) charging mode [9]. Notably, during the charging process, the equivalent resistance of the battery undergoes significant changes. This requires the charging current (voltage) in the CC (CV) charging mode to be independent of load resistance. In addition, zero phase angle (ZPA) characteristics, which are integral to the efficient operation of the system, should be maintained throughout the charging cycle.

Currently, most reported technologies for realizing CC and CV charging in WPT systems can be classified into two types. The first type involves using CC and CV control methods based on compensation topology invariance, such as phase shifting control, DC-DC converter, and frequency modulation control, as proposed in [10–13]. The output current or voltage is controlled by adjusting the phase-shifting angle of the inverter or the rectifier [10]. However, this method is prone to causing hard on/off for the switch and increasing power loss. Although these problems can be avoided by introducing a DC-DC converter to regulate the output voltage and current on the transmitter and receiver sides [11], the introduction of passive components could lead to additional power loss and create issues related to volume and weight. Notably, CC and CV charging modes can be realized at two system frequencies when considering a fixed compensation topology, indicating frequency modulation control. For instance, when using a topology with capacitors in the series compensated primary and secondary sides (S-S topology), the system produces not only the characteristic CC output at the resonance frequency point, but also two CV output frequencies that are either higher or lower than the resonant frequency [12]. These same characteristics were observed for LCC-LCC topology [13]. However, the variation in mutual inductance between the transmitter coil and the receiver coil may affect the CC or CV operating frequency points of the system, in turn reducing its reliability. Moreover, communication is indispensable to the above methods. The delay or interruption in communication inevitably influences the output characteristics of the WPT system.

As for the second type, CC and CV charging modes can also be achieved by switching two different compensation topologies [9, 14–16]. When two different topologies exhibit CC and CV output characteristics at the same resonant frequency, the changing over of the charging mode is realized by switching one or more of the AC switches. For instance, Qu et al. [9] proposed hybrid topologies using either series-series (SS) and series-parallel (PS) compensation or parallel-series (PS) and parallel-parallel (PP) compensation to realize CC and CV charging modes, achieving conversion from CC to CV charging mode using AC switches. This topology involved one inductance and three additional AC switches. Furthermore, a hybrid inductive power transfer battery charger combining LCC-LCC and LCC-S topologies was presented in [17]. The LCC-LCC compensation topology was utilized to provide the configurable CC output, while the LCC-S compensation network was employed to realize the configurable CV output. Two extra power switches were used to change the operation mode. However, in the case of the charging mode transition from CC output to CV output, an inductor on the receiver side became redundant, causing additional power loss.

This paper proposes a novel hybrid topology consisting of a reconfigurable rectifier. An AC switch is employed to facilitate the switching of the rectifier operating modes and compensation topologies, and also to enable the system to switch from CC charging mode to CV charging mode. At the same time, the proposed method does not require communication between the transmitter and receiver sides, which is beneficial for the simple and reliable operation of WPT systems.

II. Theoretical Analysis

1. A Hybrid WPT Topology with a Reconfigurable Rectifier

Fig. 1 presents the proposed WPT system with a reconfigurable rectifier, where S₁–S₄ denote four MOSFETs and D₁–D₄ are four diodes. L₁ and L₂ represent the inductance of the transmitter and receiver coils, respectively. Furthermore, C₁ is the compensation capacitor on the transmitter side, while C₂ and L₃ are the compensation capacitor and compensation inductance on the receiver side, respectively. C_r refers to the filter capacitor on the rectifier while R_L is the load resistance.

Fig. 1

The proposed WPT system with a reconfigurable rectifier.

In addition, S is an AC switch composed of two anti-series connected MOSFETs on the receiver side, as shown in Fig. 2.

Fig. 2

Diagram of AC switch S.

2. CC Charging Mode

The switch S not only controls the operating mode of the rectifier but also dominates the charging mode of the WPT system. The simplified circuit diagram of the system when S is turned off is depicted in Fig. 3, where the rectifier of the proposed WPT system works in the half-bridge rectifier (HBR) mode. The dotted line indicates the direction of the current flow on the receiver side. In HBR mode, C₁ and C₂ together form an S-S compensation network for the WPT system. Notably, the equivalent AC load of the HBR and battery load can be expressed as follows [18]:

Fig. 3

HBR mode on the receiver side.

(1) RAC_H=2π2RL.

Furthermore, according to the fundamental harmonic approximation method, the equivalent AC circuit of the HBR in Fig. 3 can be structured as shown in Fig. 4.

Fig. 4

Equivalent AC circuit of the HBR mode.

Subsequently, drawing on Kirchhoff voltage law (KVL), the following equation was obtained:

(2) {U1=(jωL1+1jωC1) I1+jωMI20=-jωMI1+(jωL2+1jωC3+RAC_H) I2.

When the operating frequency of the system becomes the resonant frequency—i.e., it is satisfied—the input current and output current can be expressed by drawing on Eq. (2), as follows:

(3) {U1=(jωL1+1jωC1) I1+jωMI20=-jωMI1+(jωL2+1jωC3+RAC_H) I2

As evident in Eq. (3), since the output current is independent of load resistance in HBR mode, CC charging can be achieved. The input impedance of the WPT system in the HBR mode can be expressed as:

(4) Zin=U1I1=(ωM)2RAC_H.

Ultimately, based on Eq. (4), the ZPA operation in the CC charging mode was obtained.

3. CV Charging Mode

A simplified circuit diagram of the system when switch S is turned on is shown in Fig. 5. Notably, the rectifier of the proposed WPT system works in full-bridge rectifier (FBR) mode.

Fig. 5

FBR mode on the receiver side.

As shown in Fig. 5, when the rectifier operates in FBR mode, it performs equivalently to two HBRs connected parallelly, with the voltage of the battery determined by the higher value of |V_ao| and |V_bo| [19]. When |V_ao| > |V_bo|, diode D₄ is clamped and D₃ is conducted. After the current flows through D₁ and the load from L₃, some of it flows through L₂ and back to L₃, while the other part flows back to L₃ from C_2. Notably, D₂ refers to the freewheeling diode, as shown in Fig. 6(a). Similarly, when |V_ao| < |V_bo|, diode D₁ is clamped and D₂ is turned off. As a result, the current flows through the load from D₄, where D₃ is the free-wheeling diode, as shown in Fig. 6(b).

Fig. 6

Two operations of the FBR mode: (a) |V_ao| > |V_bo| and (b) |V_ao| < |V_bo|.

Subsequently, the equivalent AC circuit of the FBR mode can be structured the same way as shown in Fig. 7.

Fig. 7

Equivalent AC circuit of the FBR mode.

In Fig. 7, R_{AC_F1} and R_{AC_F2} vary with changes in the load voltage. The relationship between R_{AC_F1}, R_{AC_F2}, and R_L can be expressed as follows [19]:

(5) RAC_F1//RAC_F2=2π2RL.

When the load voltage exceeds a certain rated value, the working mode of the system switches from Fig. 6(a)–6(b), with the equivalent AC circuit being the one depicted in Fig. 4. This may also be regarded as the branch disconnection of I₃ in Fig. 7, the detailed discussion for which has been presented in following section. Using KVL, the following equation was obtained:

(6) {U1=(jωL1+1jωC1) I1+jωMI20=-jωMI1+(jωL2+1jωC3+RAC_F2) I2+1jωC2I30=1jωC3I2+(jωL3+1jωC2+RAC_F1) I3.

When the system operating frequency becomes the resonant frequency, i.e., jωL₁+1/jωC₁=jωL₂+1/jωC₂=0 and L₂ = L₃ are established, the current flow through each inductor can be formulated by drawing on Eq. (6), as follows:

(7) {I1=(ωL2)2+RAC_F1RAC_F2RAC_F1(ωM)2U1I2=1jωMU1I3=-L2RAC_F1MU1.

Furthermore, the voltage of the equivalent AC load of the FBR and battery load V_ao and V_bo can be calculated as follows:

(8) {Vao=-L2MU1Vbo=RAC_F2jωMU1.

Combining Eq. (8), it is evident that when |V_ao| > |V_bo|, CV charging mode can be achieved. In CV charging mode, the voltage across C_r does not change suddenly. Moreover, the input impedance in the FBR mode can be expressed as follows:

(9) Zin=U1I1=RAC_F1(ωM)2(ωL2)2+RAC_F1RAC_F2.

Furthermore, this study calculated the ZPA operations in CV charging mode using Eq. (9). However, as evident from Eq. (8), an increase in load resistance would cause |V_bo| > |V_ao| to no longer hold true. In other words, when the load resistance value exceeds a critical value, the system would no longer operate in CV mode. In Eq. (8), assuming that V_ao is equal to V_bo, then R_{AC_F2}=ωL₂ can be obtained. Moreover, when |V_bo| > |V_ao|, the diodes D₁ and D₂ would be clamped. Subsequently, the state of the circuit will return to the FBR mode and its equivalent circuit model, as shown in Fig. 4, where R_{AC_F2}=R_{AC_H}. Drawing on the above discussion, the following conditions can be obtained using the following equation:

(10) RL_cr=π2ωL22.

Notably, for WPT systems that operate in the standard frequency band for commercial use, the critical resistance R_{L_cr} usually exceeds several hundred ohms—a value that is difficult to reach when considering the internal resistance of the battery. Thus, a |V_bo| > |V_ao| situation is not considered in the following discussions in this article.

A control flowchart of the proposed WPT system for CC/CV charging is presented in Fig. 8. When the process of charging begins, switch S remains turned off, during which the rectifier works in HBR mode and the WPT system works in CC charging mode. Subsequently, when the load voltage U_out is detected to be greater than the preset voltage U_CV, switch S is turned on. As a result, the WPT system starts working in CV charging mode using FBR mode. Lastly, the charging process concludes when the load current is finally less than the preset current I_lim.

Fig. 8

Switching control flow chart of the proposed method.

III. Experimental Validation

To validate the feasibility and superiority of the theoretical analysis demonstrated in the previous section, a 48 V/0.7 A experimental prototype was designed and then fabricated, as shown in Fig. 9. The four MOSFETs are the GS66508B located on the inverter, while the four diodes used on the reconfigurable rectifier are IV1D12030U3. Notably, the variation in load resistance was simulated using an electronics load. Fig. 10 shows a photograph of the magnetic coupler of the transmitter coil, along with a ferrite black plate (PC40). Furthermore, the specific parameters are represented in Table 1. To ensure the accuracy of the compensation inductance L₃, it was given the same winding as the receiver coil and was placed at a considerable distance from the magnetic coupler.

Fig. 9

Experimental prototype of the proposed WPT system.

Fig. 10

Transmitter coil of the proposed WPT system.

Table 1

System specifications and their designed parameters

At the beginning of the charging process, the system works in the CC charging mode. When battery resistance reaches 50% load condition, the system switches from CC charging mode to CV charging mode, and continues to maintain the CV charging mode until the end of the charging process.

As shown in Fig. 11, the fluctuation percentage of the charging current in CC charging mode is less than 1.6% (0.717–0.705 A) while that of the charging voltage in CV charging mode is below 6.2% (48.21–51.42 V). This indicates that the proposed WPT system with a reconfigurable rectifier maintains agreeable CC/CV output characteristics throughout the charging process. Fig. 12 traces the waveforms of the input voltage U₁ and input current I₁ at different load conditions in the CC charging mode.

Fig. 11

Charging process of the proposed WPT system.

Fig. 12

Waveforms of input voltage U₁ and input current I₁ in CC charging mode: (a) 25% load condition and (b) 50% load condition.

Fig. 12(a) and 12(b) show that changes in the load condition do not affect the input impedance angle between input voltage U₁ and input current I₁. Furthermore, the ZPA characteristics in the CC charging mode were verified. Fig. 13 presents the waveforms for load voltage U_out and load current I_out in the CC charging mode at the moment when the load resistance is switched from a 25% load condition to a 50% load condition.

Fig. 13

Waveforms of load voltage and current at the moment of load resistance switching in CC mode.

It is evident that at the moment of load resistance switching, the load current re-attains its original value after a brief decrease, while the load voltage increases by half. A similar analysis was conducted for the CV charging mode, as shown in Figs. 14 and 15.

Fig. 14

Waveforms of input voltage U₁ and input current I₁ in CC charging mode: (a) 50% load condition and (b) 100% load condition.

Fig. 15

Waveforms of load voltage and current at the moment of load resistance switching in CV mode.

Furthermore, the ZPA characteristic in the CV charging mode was verified, the results of which are shown in Fig. 14. Notably, a change in load condition did not influence the input impedance angle between input voltage U₁ and input current I₁. Fig. 15 depicts the load voltage and load current waveforms in the CV charging mode on switching the load resistance from a 50% load condition to a 100% load condition. It is observed that the load current decreases by half, while the load voltage undergoes a slight change, only to subsequently return to a constant value.

As observed in Fig. 16, the power transfer efficiency of the entire charging process was successfully maintained above 86.23%. In the CC charging mode, the power transfer efficiency increased gradually with an increase in the load condition, with the maximum power transfer efficiency being 95.33%. Similar to the CC charging mode, the power transfer efficiency in the CV charging mode climbed from 88.76% to 94.28% with an increase in the load condition.

Fig. 16

Power transfer efficiency of the charging process.

Table 2 provides an overview of a few of the typical methods currently used for implementing CC/CV charging modes, including their performance [9, 13, 20]. Compared to previous related studies, the method proposed in this paper offers marked improvements, such as fewer passive components and two charging modes achieved using only one AC switch at a fixed system frequency, which effectively reduced the difficulty of system control as well as system losses caused by the additional introduction of devices.

Table 2

Comparison of the current work with previous related works

Furthermore, all the controls were based on the secondary side, while the communication module could be avoided entirely. Hence, the proposed method is extremely suitable for scenarios in which communication between the primary and secondary sides is unstable, such as underwater or mine applications.

IV. Conclusion

This paper proposes a hybrid topology equipped with a reconfigurable rectifier for WPT systems. An AC switch is adapted to realize the change from the CC charging mode to the CV charging mode. When the switch is turned off, the rectifier operates in the HBR mode in conjunction with the S-S type compensation at the pre-stage, thus implementing the CC charging mode. In contrast, when the switch is turned on, the rectifier operates in the FBR mode, due to which the compensation structure of the system shifts to attain the S-LCL type, which then exhibits CV output characteristics. Notably, the ZPA condition could be maintained in both charging modes. In addition, the communication between the transmitter side and the receiver side could be eliminated in the proposed method. Furthermore, the experimental results show that fluctuations in charging current due to load condition changes in the CC charging mode can be maintained below 1.6%. For the CV charging mode, the fluctuations in charging voltage due to load resistance variation were less than 6.2%. Furthermore, during the charging cycle, power transfer efficiency was maintained above 86.23% throughout the process, with the maximum power transfer efficiency of the system reaching 95.33%. Overall, both the theoretical analysis and experimentation validate that the proposed hybrid WPT topology equipped with a reconfigurable rectifier exhibits agreeable CC/CV output characteristics throughout the charging process.

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